A vehicle lamp can generally switch between a low beam and a high beam. The low beam provides a predetermined illumination for a nearby area and has light distribution designed to not give glare to an oncoming vehicle or a preceding vehicle. The low beam is mainly used when driving in urban areas. On the other hand, the high beam provides a bright illumination for a front wide area and a distant area. The high beam is mainly used when driving at high speed on a road with few oncoming vehicles or preceding vehicles. Therefore, although the high beam gives better visibility to a driver than the low beam does, the high beam would give glare to a driver of a preceding vehicle or a pedestrian in front of the vehicle (for example, JP-A-2015-153657).
In recent years, Adaptive Driving Beam (ADB) technique has been proposed which dynamically and adaptively controls a light distribution pattern of a high beam based on conditions surrounding the vehicle. The ADB technique reduces glare to a vehicle or a pedestrian by detecting presence of a preceding vehicle, an oncoming vehicle or a pedestrian in front of the vehicle and reducing light of an area corresponding to the detected vehicle or pedestrian.
A vehicle lamp with an ADB function will be described. FIGS. 1A and 1B are block diagrams of a vehicle lamp with the ADB function according to comparative technique. The comparative technique should not be recognized as a prior art.
Referring to FIG. 1A, a vehicle lamp 1R includes a light source 2 and a lighting circuit 20R. A high beam illumination area of the ADB is divided into N sub-areas (N is a natural number equal to or more than 2). The light source 2 includes a plurality of light emitting elements 3_1 to 3_N corresponding to the N sub-areas, respectively. Each light emitting element 3 is a semiconductor device, such as a Light Emitting Diode (LED) or a Laser Diode (LD), and is disposed to illuminate the corresponding sub-area.
The lighting circuit 20R receives a power supply voltage VBAT from a battery 4 and changes a light distribution of the high beam by individually controlling ON (lighting-on) and OFF (lighting-off) of each of the light emitting elements 3_1 to 3_N. Further, the lighting circuit 20R adjusts an effective luminance by controlling a current ILAMP flowing into the light emitting element 3 by Pulse Width Modulation (PWM) at high frequency.
The lighting circuit 20R includes a boost-buck converter 22, a bypass switch circuit 24, and a light distribution controller 26. The boost-buck converter 22 is a constant current converter which generates an output current ILAMP stabilized at a target value IREF to be supplied to the light source 2.
The bypass switch circuit 24 includes a plurality of bypass switches 28_1 to 28_N corresponding to the plurality of light emitting elements 3_1 to 3_N. Each bypass switch 28_i is connected in parallel with the corresponding light emitting element 3. When the bypass switch 28_i is off, the driving current ILAMP flows into the corresponding light emitting element 3_i such that the light emitting element 3_i lights on. When the bypass switch 28_i is on, the driving current ILAMP flows into the bypass switch 28_i such that the corresponding light emitting element 3_i lights off.
The light distribution controller 26 controls ON/OFF of the plurality of bypass switches 28_1 to 28_N based on a light distribution pattern. Further, the light distribution controller 26 controls the plurality of bypass switches 28_1 to 28_N individually by PWM to perform PWM dimming of the plurality of light emitting elements 3_1 to 3_N.
Assuming the driving current ILAMP flows into M light emitting elements among the plurality of light emitting elements 3_1 to 3_N (0≤M≤N), a voltage between both ends of the light source 2, i.e., an output voltage VOUT of the boost-buck converter 22, is M×VF. Here, for ease of understanding, a forward voltage VF of the light emitting element 3 is assumed to be uniform. Therefore, the output voltage VOUT of the boost-buck converter 22 varies by time based on the combination of ON and OFF of the plurality of bypass switches 28_1 to 28_N.
As described above, the boost-buck converter 22 can be regarded as a constant current source which generates the constant driving current ILAMP. It is noted that the boost-buck converter 22 does not actively change the output voltage VOUT. The output voltage VOUT changes as a result of dynamical changing of the combined impedance of the light source 2 and the bypass switch circuit 24, i.e., the load impedance of the boost-buck converter 22.
Referring to FIG. 1B, a lighting circuit 20S includes a boost converter 30 and a buck converter 32 connected in series in place of the boost-buck converter 22 of FIG. 1A. When VF=5 V and N=12, the voltage between both ends of the light source 2 varies dynamically from 0 to 60V. The boost converter 30 is a constant voltage converter which stabilizes an output direct current voltage VDC at a voltage level higher than the maximum value 60V of the voltage between both ends of the light source 2. The buck converter 32 has a constant current output similar to that of the boost-buck converter 22 of FIG. 1A and stabilizes the current ILAMP of the light source 2 at a predetermined target value.
After examining the lighting circuit 20S of FIG. 1B, the inventors have recognized the following problems. That is, since the frequency of the PWM dimming performed by the bypass switch circuit 24 is several hundreds of Hz, the load impedance of the buck converter 32 also changes at several hundreds of Hz. In order to realize such high-speed responsiveness, it is necessary to perform hysteresis control (Bang-Bang control) in the buck converter 32. FIG. 2 is a circuit diagram of the buck converter 32 with the hysteresis control.
The buck converter 32 includes an output circuit 40 and a hysteresis controller 50. The output circuit 40 includes an input capacitor C1, a switching transistor M1, a rectifier diode D1, an inductor L1, and a current sense resistor RCS.
In the hysteresis control, an upper limit current IUPPER and a bottom limit current IBOTTOM are defined in proximity to a target value IREF of the driving current ILAMP which is a control object. The switching transistor turns off when the driving current ILAMP (coil current IL) reaches the upper limit current IUPPER, and the switching transistor turns on when the driving current ILAMP falls to the bottom limit current IBOTTOM, and this cycle is repeated.
The hysteresis controller 50 includes a current detection circuit 52, a hysteresis comparator 54, and a driver 56. The current sense resistor RCS is provided on a path of the driving current ILAMP. A voltage drop proportional to the driving current ILAMP is generated between both ends of the current sense resistor RCS. The current detection circuit 52 generates a current detection signal VCS corresponding to the voltage drop of the current sense resistor RCS. The hysteresis comparator 54 compares the current detection signal VCS with an upper threshold VTHH corresponding to the upper limit current IUPPER and a bottom threshold VTHL corresponding to the bottom limit current IBOTTOM, and generates a control pulse SCNT corresponding to the comparison results. The driver 56 drives the switching transistor M1 according to the control pulse SCNT.
FIG. 3 is an operation waveform diagram of the buck converter 32 of FIG. 2. In a section where the control pulse SCNT is at an ON level (for example, a high level), the switching transistor M1 is on, and in a section where the control pulse SCNT is at an OFF level (for example, a low level), the switching transistor M1 is off. When the switching transistor M1 is on, a voltage between both ends of the inductor L1 is VIN−VOUT. Therefore, the coil current IL flowing into the inductor L1 (i.e., the driving current ILAMP) rises with a slope of (VIN−VOUT)/L1. When the switching transistor M1 is off, a voltage between both ends of the inductor L1 is −VOUT. Therefore, the coil current IL (i.e., the driving current ILAMP) falls with a slope of −VOUT/L1.
The ON time TON and OFF time TOFF of the switching transistor M1 are given by equations (1) and (2).TON=ΔI/{(VIN−VOUT)/L1}  (1)TOFF=ΔI/(VOUT/L1)  (2)
ΔI is a hysteresis width (ripple width) of the coil current IL, that is, the difference between a peak value IUPPER a and a bottom value IBOTTOM. ΔI is proportional to the difference ΔV between the upper threshold signal VTHH and the bottom threshold signal VTHL as shown in the following equation.ΔI=ΔV/RCS 
In the vehicle lamp 1S of FIG. 1B, although an input voltage VIN (VDC) of the buck converter 32 remains constant, the output voltage VOUT fluctuates dynamically with the PWM control of the bypass switch circuit 24. The switching period TON+TOFF of the switching transistor M1, i.e. switching frequency fSW, fluctuates with the output voltage VOUT, which makes it difficult to deal with electromagnetic noise.